Position sensorless control algorithm for AC machine

ABSTRACT

A control system for an electric motor having a stator and rotor including an inverter for providing power to the electric motor, a controller for controlling the inverter, a low speed control block to estimate the rotor angular position using stator current components operating in the controller, a high speed control block to estimate the rotor angular position using stator current components and stator flux position operating in the controller, a transition switch in the controller to vary operation between the low speed control block and the high speed control block, and where the inverter is controlled by six step operation.

TECHNICAL FIELD

The present invention relates to the control of electric motors. Morespecifically, the present invention relates to a method and apparatusfor position sensorless control of an electric motor.

BACKGROUND OF THE INVENTION

Traditional motor control systems normally include a feedback device orposition sensor such as a resolver or encoder to provide speed andposition information for a motor. Feedback devices and associatedinterface circuits increase the costs of a motor control system, andthese costs may become prohibitive in high volume applications such asautomotive applications. Additionally, a position sensor and itsassociated wiring harness increase the complexity and assembly time ofan electric drive system in a vehicle.

Electric vehicles powered by fuel cells, batteries and hybrid systemsthat include electric motors are becoming more common in the automotivemarket. As production volumes for electric vehicles increase, the costof feedback devices and associated interface circuits will becomesignificant. Automakers are under intense market pressure to cut costsand reduce the number of parts for a vehicle. The removal of a feedbackdevice for an electric motor control system will lead to significantcost reductions for an electric vehicle.

Hybrid electric and electric vehicles today utilize numerous electricmotor control technologies such as the vector control of electricmotors. A vector motor control scheme is a computationally intensivemotor control scheme that maps the phase voltages/currents of athree-phase motor into a two-axis coordinate system. The structure usedto excite an electric motor using a vector control scheme is a typicalthree-phase power source inverter including six power transistors thatshape the output voltage to an electric motor. Vector control requiresrotor position information, which is normally obtained via a feedbackdevice or position sensor. The objective of the position sensorlesscontrol is to obtain the rotor position information utilizingelectromagnetic characteristics of an AC machine, eliminating theposition sensor and its associated interface circuits.

SUMMARY OF THE INVENTION

The present invention is a method and apparatus for a sensorless controlsystem used in electric and hybrid electric vehicle powertrainapplications. The sensorless motor control system of the presentinvention includes a low-speed angular position estimation method, aninitial rotor polarity detection method, a transition algorithm betweenlow and high speed methods, a modified Gopinath observer, a fieldweakening method, and/or six step operation.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a control system in the present invention;

FIG. 2 is a vector diagram illustrating possible orientation frames forcontrol of the present invention;

FIG. 3 is a block diagram of a modified Gopinath observer used in thepresent invention;

FIG. 4 is a block diagram of a controller used for six step operation inthe present invention; and

FIG. 5 is a state flow diagram for transition into and out of six stepoperation.

DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 1 is a diagrammatic drawing of a preferred embodiment of a controlsystem 10 of the present invention. The control system 10 is illustratedas a sequence of block diagrams that represent software executed in acontroller, microprocessor, or similar device to control an electricmotor 12. In the preferred embodiment of the present invention, thecontroller is a vehicle powertrain controller controlling the electricmotor 12, but any other motor control application is considered withinthe scope of the present invention. The electric motor may comprisemotor technologies such as AC machines, synchronous reluctance motors,induction motors and interior permanent magnet motors, but is notlimited to such. The input to the control system is a torque commandT_(e) generated by the vehicle controller. The torque command T_(e) isprocessed by a optimum torque/amperage calculation block 14 to generatea corresponding stator current command I_(s) and current angle command βrequired to develop the desired electromagnetic torque in the motor 12.A field weakening stator flux λ_(fw) is generated at field weakeningblock 15 based on the measured DC link voltage V_(dc) and rotor angularvelocity ω_(r). Block 16 will modify the commands λ and δ if the statorflux command λ exceeds the field weakening stator flux command λ_(fw).

The stator current command I_(s) and current angle command β generatedat bock 14 are passed to a stator flux and torque angle calculationblock 16. Block 16 processes the commanded stator current I_(s) andcurrent angle command β and decomposes it into the stator flux command λand torque angle command δ to provide the maximum torque for the givenstator current amplitude.

Summing junction 18 subtract the feedback stator flux λ_(fb) from thestator flux command λ to generate an error. Summing junction 20subtracts the feedback torque angle δ_(fb) from the torque angle commandδ_(fb) to generate an error. The error generated by summing junction 18is processed by Proportional Integral (PI) control block 22 to generatea control output. The error generated by summing junction 20 isprocessed at multiplication block 24 where it is multiplied by thefeedback stator flux λ_(fb). The output of block 24 is processed atproportional control block 26.

The output of PI control block 22 is summed at summing junction 28 withthe stator resistance voltage drop decoupling term R_(s)I_(f) to producethe f-axis voltage command V_(f). The output of proportional controlblock 26 is summed at summing junction 30 with the τ-axis decouplingterm (stator resistance voltage drop plus speed voltage)ω_(r)λ_(fb)+R_(s)i_(τ) to produce the τ-axis voltage coomand V_(τ).V_(f) and V_(τ) are processed at the rotating to stationary frametransformation block 32 using the estimated stator flux angular positionθ_(f) to convert the stator flux reference frame voltage commands V_(f)and V_(τ) to the stationary frame voltage commands V_(α1) and V_(β1).High frequency injection signals V_(α) _(—) _(inj) and V_(β) _(—) _(inj)are added at summing junctions 34 and 36 to the stationary frame voltagecommands V_(α1) and V_(β1) to produce final voltage commands V_(α) andV_(β) that generate the actual phase voltage commands applied to theelectric motor 12. The voltage source inverter 38 processes the finalvoltage commands V_(α) and V_(β) using a two-phase to three-phasetransformation to generate the actual three-phase voltages to be appliedto the motor 12.

The phase currents are measured and processed by a three-phase totwo-phase transformation block 40. The outputs of the block 40 arestationary frame currents I_(α) and I_(β). A stationary to rotatingframe transformation block 42 uses the stationary frame currents I_(α)and I_(β) and the estimated rotor angular position θ_(f) to generatestator flux reference frame feedback currents I_(f) and I_(τ).

The present invention further includes position sensorless control ofthe machine that includes: a low speed rotor angular position estimationmethod/observer at block 44; an initial rotor polarity detection methodat block 46; a high speed rotor angular position estimationmethod/observer at block 48; a modified Gopinath observer block 50; anda transition switch 54 to seamlessly merge the low and high speedestimation methods.

Block 44 of FIG. 1 represents the low speed estimation method of thepresent invention. The low speed method 44 uses stationary referenceframe stator current components I_(α) and I_(β) to estimate the rotorangular position θ_(r) _(—) _(low). Similarly, the high speed method 48uses stationary reference frame stator current components I_(α) andI_(β) and estimated stator flux position θ_(f) to estimate the rotorangular position θ_(r) _(—) _(high). The switch 54 selects appropriaterotor angular position θ_(r) based upon estimated rotor speed.

The modified Gopinath observer 50 processes θ_(r), the stationary framevoltages V_(α) and V_(β), and stationary reference frame currents I_(α)and I_(β). The modified Gopinath observer 50 computes the estimatedstator flux angle θ_(f), the feedback stator flux λ_(fb), and thefeedback torque angle flux δ_(fb).

In the preferred embodiment of the present invention, the d-axisinductance will be higher than the q-axis inductance, and the machinemagnet north pole is oriented in the (−) q-axis direction. However, theproposed control scheme will still be valid if the machine q-axisinductance is higher than the d-axis inductance. FIG. 2 is a vectordiagram showing possible orientation frames for control. The α and βaxes are used for the stationary reference frame control. In thestationary reference frame control variables are the AC time varyingsignals. It is preferable to use a rotating reference frame for control,where the control variables are DC quantities. The synchronous referenceframe (rotor-oriented reference frame, or d-q frame) and the stator fluxreference frame (f-τ reference frame) are both rotating reference frameswith DC control variables in steady state.

For a highly saturated machine, the d-q frame voltage equations havebi-directional cross coupling terms, which can limit synchronous currentcontroller bandwidth. Equation (1) below shows the stator voltageequations in the d-q reference frame illustrating the cross-couplingeffects. $\begin{matrix}\begin{matrix}{v_{ds}^{e} = {{r_{s}i_{ds}^{e}} + {L_{d}^{\prime}\frac{\mathbb{d}i_{ds}^{e}}{\mathbb{d}t}} + {i_{ds}^{e}\frac{\partial L_{d}}{\partial i_{qs}^{e}}\frac{\mathbb{d}i_{qs}^{e}}{\mathbb{d}t}} - {\omega_{r}\lambda_{qs}^{e}}}} \\{v_{qs}^{e} = {{r_{s}i_{qs}^{e}} + {L_{q}^{\prime}\frac{\mathbb{d}i_{qs}^{e}}{\mathbb{d}t}} + {i_{qs}^{e}\frac{\partial L_{q}}{\partial i_{ds}^{e}}\frac{\mathbb{d}i_{ds}^{e}}{\mathbb{d}t}} + {\omega_{r}\lambda_{ds}^{e}}}}\end{matrix} & (1)\end{matrix}$

In Equation (1), the last two terms in each voltage equation arecross-coupling terms. Since the d-axis inductance is much larger thanthe q-axis inductance, the time constant in the d-axis is much longerthan that in the q-axis. Any disturbance introduced into the d-axisvoltage equation due to the cross-coupling terms will have minimaleffect on the d-axis current regulation due to the long time constant.However, the disturbance introduced by the cross-coupling terms into theq-axis voltage equation will have a significant effect on the q-axiscurrent regulation. As a result, attempting to increase the currentregulator bandwidth will result in unstable operation. To overcome theselimitations, it is desirable to change the controller reference frame tothe stator flux reference frame (f-τ reference frame). The machineequations in the stator flux reference frame can be described as shownin Equations (2) and (3) below: $\begin{matrix}{\frac{\mathbb{d}\lambda}{\mathbb{d}t} = {v_{f} - {r_{s}i_{f}}}} & (2) \\{{\lambda\quad\frac{\mathbb{d}\delta}{\mathbb{d}t}} = {v_{\tau} - {r_{s}i_{\tau}} - {\omega_{r}\lambda}}} & (3)\end{matrix}$From Equation (2), it can be seen there is no cross-coupling from the τaxis into the f-axis. However, there is uni-directional coupling fromthe f-axis into the τ-axis. The uni-directional cross-coupling is easyto decouple in the control. For the above reasons, it can be seen thatthe stator flux oriented control is more suited for this type of machinecompared to the rotor flux oriented control.

The modified Gopinath observer 50 shown in FIG. 3 is used to estimatethe stator flux angle θ_(f), the feedback stator flux λ_(fb), and thefeedback torque angle δ_(fb). The stationary frame currents I_(α) andI_(β) are input to the observer. Stationary to synchronous referenceframe transformation module 60 transforms the stationary frame currentsto the synchronous reference frame using rotor angular position θ_(r).The machine current model 62 calculates the stator flux of the machinein the synchronous reference frame. Synchronous to stationary referenceframe transformation module 64 transforms the synchronous frame statorflux to the stationary reference frame using rotor angular positionθ_(r).

Stator resistance gain module 66 and summer 68 are used along with thestationary reference frame voltages V_(α) and V_(β) and currents I_(α)and I_(β) to calculate the stationary frame back EMF. The integrator 70is used to integrate the back EMF to calculate the stator flux basedupon the voltage model.

The current model is more accurate at lower speeds, while the voltagemodel based calculation is more accurate at higher speeds. Therefore,blocks 72, 74, and 76 are used to smoothly transition the stator fluxcalculation from current model to voltage model based upon rotor speed.Equation (4) describes how blocks 72, 74, and 76 result in smoothtransition between the current model flux estimate λ_(αβ-CM) and thevoltage model flux estimate λ_(αβ-VM) as a function of the electricalfrequency ω_(θ). The observer characteristic function F(s) is also shownin Equation (4). The setting of module 72 PI gains is shown in Equation(5). Block 76 assures optimal transition trajectory between the currentmodel and voltage model estimated stator flux vectors. Module 80 is usedto calculate the stator flux angular position Of using an arctangentfunction. $\begin{matrix}\begin{matrix}{\lambda_{\alpha\quad\beta} = {{{F(s)} \cdot \lambda_{{\alpha\quad\beta} - {VM}}} + {\left\lbrack {1 - {F(s)}} \right\rbrack \cdot \lambda_{{\alpha\quad\beta} - {CM}}}}} \\{{F(s)} = {\frac{s^{2}}{s^{2} + {K_{p}s} + K_{i}} \cdot {\mathbb{e}}^{{- j}\quad\alpha}}} \\{\alpha = {\pi - {\tan^{- 1}\left( \frac{K_{p}\omega_{e}}{K_{i} - \omega_{e}^{2}} \right)}}}\end{matrix} & (4)\end{matrix}$

-   -   where    -   K_(p)√{square root over (2)}·ω_(c)    -   K_(i)=ω_(c) ²  (5)    -   ω_(c)=transition frequency

The modified Gopinath observer 50 is used to estimate the stator fluxangle θ_(f) at all speeds. The appropriate input θ_(r) is automaticallyselected by the transition switch 54 depending on the rotor speed.

Field-weakening operation is achieved utilizing module 15. Equation (6)is used to calculate the field-weakening stator flux command based uponDC link voltage and rotor speed. $\begin{matrix}{\lambda_{fw} = \frac{V_{D\quad C} \cdot K_{fw}}{\sqrt{3}\omega_{r}}} & (6)\end{matrix}$Under all operating conditions λ_(fw) is being compared to λ. The lowerflux command is used by the controller as the final flux reference. Ifthe field-weakening stator flux command λ_(fw) is selected, then commandcalculation module 16 recalculates optimum torque angle command δ basedupon the new flux command λ_(fw) and torque command T_(e).

The diagram shown in FIG. 4 illustrates the proposed control that allowsfor torque regulation during six-step operation. During six-stepoperation, the voltage applied to the stator is fixed. Hence, there isonly one degree of freedom in the controller. Torque is controlled byregulating the torque angle δ, which in turn controls the voltage angleα with respect to the d-axis of the synchronous reference frame. FIG. 5shows a state flow diagram that describes the transition into and out ofsix-step operation.

Referring to FIG. 4, the switch 100 transitions between normal operationand six-step operation. During normal operation, the diagram in FIG. 1generates the stationary frame voltage commands V_(α1) and V_(β1). Whenthe variable Flag_six becomes true, the voltage commands V_(α1) andV_(β1) are supplied by the six-step control module 102.

The six-step control module 102 regulates the torque angle δ, which inturn controls the voltage angle applied to the machine. Torque anglecommand δ is compared to torque angle feedback δ_(fb) using summingjunction 104, whose output is fed to PI regulator 106. The PI regulatorinitial state is set to provide seamless transition into and out ofsix-step operation. The feedforward voltage angle calculation module 108calculates the feedforward voltage angle α_(ff) for faster dynamicperformance. The summer 110 adds the PI regulator output and thefeedforward voltage angle α_(ff) to generate the final voltage angle α.The voltage angle α is added to the rotor angular position θ_(r) usingsumming junction 112 to produce the stationary frame voltage angle.Block 114 utilizes the output of summing junction 112 and the maximumavailable voltage (six-step voltage) to generate the command voltagesV_(α1) and V_(β1).

FIG. 5 details the state flow diagram describing the setting oftransition flag Flag_six in FIG. 4. The entire flow diagram is executedevery sample period. Decision block 120 compares the actual rotor speedω_(r) with a predefined minimum threshold speed ω_(rth). If the rotorspeed is less than the predefined minimum threshold speed, Flag_six isset to zero (normal stator flux oriented control described in FIG. 1) atblock 122. Otherwise, decision block 124 is used to compare the appliedstator voltage V_(m) to a predefined maximum threshold voltage V_(th).If the applied stator voltage is less than the V_(th), then Flag_six isset to zero at block 122. Otherwise, Flag_six is set to 1 (six-stepoperation) at block 126. The control remains in this mode of operationuntil the condition of decision block 130 is evaluated true. Block 130detects the condition where the estimated stator flux exceeds thecommanded stator flux. If TRUE, there is sufficient voltage available toexit six-step operation and return to normal stator flux orientedcontrol.

It is to be understood that the invention is not limited to the exactconstruction illustrated and described above, but that various changesand modifications may be made without departing from the spirit andscope of the invention as defined in the following claims.

1. A control system for an electric motor having a stator and rotorcomprising: an inverter for providing power to the electric motor; acontroller for controlling said inverter; a low speed control block toestimate the rotor angular position using stator current componentsoperating in said controller; a high speed control block to estimate therotor angular position using stator current components and stator fluxposition operating in said controller; a transition switch in saidcontroller to vary operation between said low speed control block andsaid high speed control block; and wherein said inverter is controlledby six step operation.
 2. The control system of claim 1 wherein saidelectric motor is an induction motor.
 3. The control system of claim 1wherein said electric motor is an interior permanent magnet motor. 4.The control system of claim 1 wherein said electric motor is asynchronous reluctance motor.
 5. The control system of claim 1 whereinsaid electric motor is a three-phase motor.
 6. The control system ofclaim 1 further comprising a Gopinath observer.
 7. The control system ofclaim 1 wherein said transition switch operates said first motor speedcontrol block below ten percent of rated machine speed.
 8. The controlsystem of claim 1 wherein said transition switch operates said secondmotor speed control block above five percent of rated machine speed. 9.A sensorless method of controlling an electric motor comprising:providing a low speed rotor angular position block operating in acontroller; providing a high speed rotor angular position blockoperating in said controller; providing an initial rotor polaritydetection block operating in said controller; transitioning between saidlow speed rotor angular position block and said high speed rotor angularposition block to determine the speed of the electric motor; andcontrolling the speed of the electric motor with six step operation. 10.The method of claim 9 further comprising operating the electric motor ina vehicle.
 11. The method of claim 9 further comprising the step ofdetermining the rotor magnet position of the electric motor in a staticstate.
 12. A powertrain for a vehicle comprising: an electric motorfunctionally coupled to a wheel of the vehicle; an inverter forproviding power to the electric motor; a controller for controlling saidinverter; a low speed control block to estimate the rotor angularposition using stator current components operating in said controller; ahigh speed control block to estimate the rotor angular position usingstator current components and stator flux position operating in saidcontroller; a transition switch in said controller to vary operationbetween said low speed control block and said second high speed controlblock; and wherein said inverter is controlled by six step operation.13. The powertrain of claim 12 wherein said electric motor is aninduction motor.
 14. The powertrain of claim 12 wherein said electricmotor is an interior permanent magnet motor.
 15. The powertrain of claim12 wherein said electric motor is a synchronous reluctance motor. 16.The powertrain of claim 12 wherein said electric motor is a three-phasemotor.
 17. The powertrain of claim 12 wherein said transition modulevaries operation between said first control module and said secondcontrol module based on the speed of the electric motor.
 18. Thepowertrain of claim 12 wherein the electric motor includes an interiorpermanent magnet rotor.